LA3600 5 Band AF equaliser

https://www.transkommunikation.ch/dateien/schaltungen/diverse_tbs_schaltungen/5%20band%20graphic%20equalizer%20using%20LA3600.pdf

Now I did have a practical wireless magazine purchased from my local news agent, but unfortunately I have some how at the moment miss-layed my copy of this month. However I did look up the data sheet for the chip design, and as I had lost my copy, I then proceeded to write up a BBC Basic code CAD program for calculating the capacitor values.

The tabled results of C_one and C_two relate to the band pass filtering of the 100K ohm variables resistor pots. Starting say from the left control resistor pot, call this pot number one, this maybe setup for the 311Hz band equalisation control of the audio signal, with C_one and C_two are 32nF and 100nF. However the next resistor pot on its right hand side, call this pot number two, here C_one and C_two are 4.7nF and 180nF for the 605Hz AF signal equalisation control setting, and so on from pots No:3 the 1081Hz setting, No:4 the 2099Hz setting and finally pot No:5, the 3116Hz control setting, and thus the appropriate C_one and C_two values as listed in the tabled values. However, should the DIY ham radio constructor wish to change the frequency settings, just run the BBC Basic CAD program and change the values of C_one and C_two for the preferred chosen equalisation audio signal requirements.

I suppose that such a circuit could be used for gaming, that is to peek the audio quality for either PC, xbox or playstation users. I am not sure how to insert such a connection, but it would be a useful introduction improvement, perhaps as the voice link of the gaming scene is I think just a voice audio link, but perhaps within the 6KHz range which is equivalent to broadcast quality AM radio. In this dual control variable resistor pots would be used, that is two pots on the same axis, usually called stereo variable resistors, one pot on the axis each for separate left and right sound channels, thus a "LA3600 device" for each channel. The reason for this is to provide a left and right hand side audio channel audio signal equalisation, but usually with head set units with a microphone, the microphone is perhaps a mono sound device. Thus an separate audio equalisation unit could be used for the microphone audio, thus a third "LA3600 device", which a fully installation would a bank of two sets on controls. A head piece stereo sound AF processor ( equalisation unit ) with the next layer the microphone processor ( equalisation unit ). I think food for thought though. Mind you as ham radio is essentially mono audio unless the band plan allows for a stereo transmission, the processor box in its either mono or stereo form could be used also for ham radio for Tx and Rx audio settings. Also there are also the CB radio users as well not to forget about, be it AM, SSB or FM transmission on CB radio.

According to the Sanyo manufactures data sheet of the LA3600 device illustrated here, the variable controls would give a +/- 10dB of a cut ( -10dB) or boost (+10dB) to the audio quality.

Apparently a 5Volt supply is adequate.

CAD program no: 1

R_one = 1200

R_two = 68000

REM 300Hz

C_one = 32E-9

C_two = 100E-9

PROC_cal_resonance

REM 600Hz

C_one = 4.7E-9

C_two = 180E-9

PROC_cal_resonance

REM 1KHz

C_one = 3.9E-9

C_two = 68E-9

PROC_cal_resonance

REM 2KHz

C_one = 2.2E-9

C_two = 32E-9

PROC_cal_resonance

REM 3KHz

C_one = 4.7E-9

C_two = 6.8E-9

PROC_cal_resonance

END

DEF PROC_cal_resonance

root = SQR(C_one*C_two*R_one*R_two)

F= 1/( 2 * PI * root)

PRINT TAB(5);"Freq = ";INT(F*1)/1;TAB(20);"C_one = ";C_one*1E9;" nF";TAB(37);"C_two = ";C_two*1E9;" nF"

ENDPROC

According to calculation, illustrated later in this article, the Q of the above manufacturer circuit diagram is Q of each filter tallys as follows:

311Hz , C1 = 32nF, C2 = 100nF, Q = 0·9

605Hz, C1 = 4·7nF, C2 = 180nF, Q = 3

1081Hz, C1 = 3·9nF, C2 = 68nF, Q = 2

2099Hz, C1 = 2·2nF, C2 = 32nF, Q = 2

3116Hz, C1 = 4·7nF, C2 = 6·8nF, Q = 0·6

For further reference, the url https://learn.sparkfun.com/tutorials/proto-pedal-example-analog-equalizer-project/the-circuit is an interesting circuit, should one wish to build a sound parametric equaliser from scratch. The next url https://sound-au.com/articles/gyrator-filters.htm is also a good reference for such circuits. Another reference that has come across is " https://circuitswiring.com/bandpass-filters-gyrators-frequency-calculator-circuit-diagram/ " .

I have also written a BBC Basic code program for gyrator circuit tuning, shown below:

CAD program No: 2

R_one = 330

Cap = 100E-9

R_two = 47000

inductor = R_one * R_two * Cap

F= 1/( 2 * PI * SQR(inductor * Cap))

PRINT TAB(5);INT(F);" Hz"

PRINT TAB(5);"R_one = ";R_one;" ohms"

PRINT TAB(5);"Cap = ";Cap*1E9;" nF"

PRINT TAB(5);"R_two = ";R_two;" ohms"

PRINT

PRINT

R_two = 470

inductor = R_one * R_two * Cap

F= 1/( 2 * PI * SQR(inductor * Cap))

PRINT TAB(5);INT(F);" Hz"

PRINT TAB(5);"R_one = ";R_one;" ohms"

PRINT TAB(5);"Cap = ";Cap*1E9;" nF"

PRINT TAB(5);"R_two = ";R_two;" ohms"

Now I admit that printing some ones circuit is perhaps not cricket, but I am about to add to the original info, so I thought I was perhaps doing no harm. From the above BBC basic tabled results, the resistor "R_two", is variable from 47K ohm down to 470 ohms. Now, the things is that this has made the bandpass filter gyrator circuit tuneable, this so, the above circuit marked "out 2", can then be adjusted to the part of the voice speech band one is interested within to either peak or reduce. However, the circuit marked "out 1", would serve nicely as a tuneable notch filter, for those signals of a tuning up transmitter on shortwave.

However. if one changes the above circuit thoughts, assuming that R2 varies from 47K ohm to minimum of 150 ohms, the frequency tuning ranges is some what wider, also assuming that R1 stay constant.

The program code changes a bit too as shown below:

CAD program no:3

R_one = 680

Cap_one = 100E-9

Cap_two = 100E-9

R_two = 47E3

inductor = R_one * R_two * Cap_one

F= 1/( 2 * PI * SQR(inductor * Cap_two))

PRINT TAB(5);"F_tune low = ";INT(F);" Hz"

PRINT

PRINT TAB(5);"R_one = ";R_one;" ohms"

PRINT TAB(5);"Cap_one = ";Cap_one*1E9;" nF"

PRINT TAB(5);"Cap_two = ";Cap_two*1E9;" nF"

PRINT TAB(5);"R_two = ";R_two;" ohms"

PRINT

PRINT

R_two = 150

inductor = R_one * R_two * Cap_one

F= 1/( 2 * PI * SQR(inductor * Cap_two))

PRINT TAB(5);"F_tune high = ";INT(F);" Hz"

PRINT

PRINT TAB(5);"R_one = ";R_one;" ohms"

PRINT TAB(5);"Cap_one = ";Cap_one*1E9;" nF"

PRINT TAB(5);"Cap_two = ";Cap_two*1E9;" nF"

PRINT TAB(5);"R_two = ";R_two;" ohms"

Now using CAD program no:1, using the BBC basic code to calculate the spot frequency of tuning for a 5 band parametric equaliser found on the internet, the circuit below uses notched filters, shown below:

The spot frequency tuning as calculated are listed on the below image.

The web design Q filter values equate as follows :-

49Hz, Q = 1

226Hz, Q = 1

889Hz, Q = 1

3930Hz, Q = 1

15651Hz, Q = 1

However, it is possible to design you own filter spec, a list below shows a range of values from 3KHz to 350Hz. I have also found out to determine the "Q" filter response of the gyrator circuit. The text box below lists the BBC Basic code, an the table a range of values.

gyrator BBC Basic coding

10 REM calculation of gyrator circuit configuration

20

30 Cone = 10E-9

40 PRINT TAB(5);"C1 = ";Cone*1E9;" nH"

50 Ctwo = 680E-9

60 PRINT TAB(5);"C2 = ";Ctwo*1E9;" nH"

70 Rone = 1500

80 PRINT TAB(5);"R1 = ";Rone;" ohms"

90 PRINT

100 Rtwo = 280 + 20000

110 FOR Rtwo = 280 TO 2000 STEP 250

120 f = 1/(2*PI * SQR(Cone * Ctwo * Rone * Rtwo))

130

140 PRINT TAB(5);"Rtwo = ";Rtwo;" f = ";f;" Hz"

150 NEXT

160 FOR Rtwo = 2500 TO 20000 STEP 2500

170 f = 1/(2*PI * SQR(Cone * Ctwo * Rone * Rtwo))

180

190 PRINT TAB(5);"Rtwo = ";Rtwo;" f = ";f;" Hz"

200 NEXT

210 PRINT

220 q = 1/2 * SQR(Cone/Ctwo)

230 PRINT TAB(5);"Q = ";q

Gyrator filter sub-section reference drawing, the above BBC basic code has been altered to match the above circuit example, here the Q = 1, mainly it seems because the ( C1 = 150nF ) value is fifteen times higher than the calculation examples below ( C1 = 10nF ) where the Q = 4. It maybe perhaps that the lower ( C1 = 10nF ), reduces the loading effect the balance variable cut or boost control has upon the filter circuit :-

Tabled values from the calculation BBC Basic Coding.

Roughly working out the bandwidth with the Q = 4, of each gyrator filter setting, the perhaps best centre frequency arrangement would be 2700Hz, 1600Hz, 900Hz, 675Hz, 375Hz, the values based on the filter circuit design "Q", the bandpass width just touch the edge of the neighbour filter response, hopefully then providing a smooth balance of the audio range from 300Hz to 3000Hz, a full width SSB filter in todays DSP radios.

Do however note, that the 20Kohm variable adjustment setting, has a pad resistor of 280 ohms in circuit to ground. The component values according to the tabled values, relate to the component values of the gyrator filter sub-section drawing. Be careful regarding the Cone and Ctwo as I am not sure which way the capacitor arrangement may be to create the Q=4. The equation may be just a little odd, it may be that depended which way the capacitors are placed according to their value, one may get a Q=4 or not so. So do check the construction under its test and commissioning conditions.

Reviewing "Section 5" of the "sound-au.com" article, the above circuit functions as an active inductor arrangement.

It seem though that the depth and boost of either the notch or bandpass versions apparently depends upon the resistor values, so I would suggest building each gyrator circuit to test the signal boost or depth of attenuation.

By following the above op-amp gyrator circuits, the "out 2" circuit with the below components values, the gyrator bandpass filter would be tune-able from 330Hz to near 6KHz.

The 33 ohms "R_two" resistor, is actually a padding 33 ohm resistor, so as the 10K pot whipper reaches to ground potential, the 33 ohm resistor will give the 10K pot circuit a minimum resistance value of 33 ohms. When designing your own discrete component parametric equaliser, it is perhaps best to use a split voltage supply source for the op-amp circuit, which can be found by using a negative voltage generator, from the filtered +5V positive supply voltage connection.

As a microphone equaliser, the +5 volts from the radio microphone connection, or a tap from the radio PSU supply unit. Some, or most radios have a balanced or settled level output of the radio's audio, which is usually not linked to the radio's main audio volume control to the radio speaker. Perhaps it may be wise to an audio transformer decoupling circuit, several available via either Amazon or Ebay, for just a few quid. I would suggest doing the same arrangement for your "Xbox" and "Play Station", not forgetting the Citizen Band Radio community "CB radio" cousins.

When designing one's own parametric sound equaliser, try not to use a variable resistor with an resistance greater than 50K ohms. Apparently, if you use a 100K ohm variable resistor, the signal current running through the pot is small enough to be effect by a mains hum. This mains hum is from the person touching the 100K ohm pot to adjust the setting position. By using a max 50K ohm pot, this problem is overcome. I discovered this snag during the build of my 40m "bitx40 radio kit".

Now with an audio parametric equaliser circuit, an audio compressor circuit is usually added to the overall design. For such a circuit implementation principle, the below circuit may be usual full.

The compressor circuit of a "T" section of the two resistors and the mosfet ( 2N7000 ) device, is found also in the bitx40 published a.g.c. circuit. In this arrangement, the agc voltage is generated from the mic audio input through an op-amp, then rectified to produce an agc control voltage. The agc voltage is made variable to give a variable audio compression to the mic audio. A variable is also suggested to set the time constant of the a.g.c. compression signal voltage, that best suits the user of the ham radio transmitter.

One may have noticed that within the compression circuit principle, there is not a feed back signal to the op-amp input. This is correct, as the circuit is arranged as above is to avoid an oscillating compression characteristic fed onto the audio from occurring. The degree of voice audio into the microphone and hence the generated audio signal, is usually set from the outset, so the above audio compressor circuit would assume an apparent level of mic audio. The voice envelope thus used as a compression AGC voltage to squeeze the dynamic range of the mic audio into the transmitter. The method being that the louder the audio from the mic, the greater the voice envelope voltage and the stronger the audio compression.

As the user distance from the mic would give a level of mic audio signal, and as there is not an automatic gain adjustment mic amplifier, the overall pre-set mic gain amplification would be not sufficient to pick up any room background noises. The audio compressor circuit would only compress any mic audio that is sufficiently load enough to attract compression. It would thus may be the case that a mic noise gate circuit may perhaps not be required.

Now the art of a de-compressor otherwise referred to as an audio expander, may be achieved as shown below:

Now, the above circuit to a point my look familiar, and that is because the mosfet ( 2N7000 ) device is at the input of the op-amp. Usually such an arrangement would have been used for a feedback control in auto level circuit amplifier. Now is so, but in this case, the technique to auto adjust an amplifiers signal gain, is feed from the voice signal envelope. According to the strength of the audio signal envelope, so the dynamic gain of the signal amplifier will alter.

By setting the expander signal de-compression to the reverse of the original signal compression, the original audio dynamic signal performance can re-obtained at the receive end, be this a TV or radio audio, or even from the audio of a movie that would be normally shown in a movie theatre. Should the sound track of the movie need to reduced by 40dB dynamic range to fit a 60dB dynamic range broadcast transmission, then the de-compressor hence the expander, could in theory recover the lost 40dB dynamic range of the movie sound track.

Mind you, the movie itself my perhaps have a compressed dynamic range sound track, to fit on the movie sound track system in the first place. If so, then after the de-compressor ( sound expander ), the original "real world sound recording", would then be perhaps recovered.

Just as a foot note, while drawing up the audio sound expander circuit, I originally put the mosfet ( 2N7000 ) in front of R1. Now if the op=amp was a unit gain, in either the compander or de-compander, then on the compander circuit the mosfet channel resistance would add to the value R1, but if the mosfet was in parallel with R1, the parallel circuit would have a reducing total resistance to R1. to infront of R1, thus creating a compander or expander amplifier circuit.

To create a compander circuit from an op-amp, the mosfet infront of R1, would normally be biased on ( lowest channel "on" resistance ) with a pull_up resistor to a reference voltage source. The sound envelope control signal would then activate another 2N7000 to pull the reference voltage of the input of the dynamic range reducing mosfet ( the mosfet in-front of R1 ), thus then increasing the mosfet channel resistance, then the op-amp acting as a compressor circuit.

The reason is that the combination of the varying "dynamic range reducing" mosfet channel "on" resistance plus the value of R1, would be greater in total than Rf, the op-amp gain feedback resistors resistance, thus creating a reducing signal gain amplifier,, the reducing gain according to the overall resistance of R1 and the varying channel "on" resistance of the inline series mosfet, the op-amp overall acting thus as a compressor circuit.

It may be that the greater the number of mosfet used, in either series or parallel, the greater the generated dynamic range of compression or expansion.

However, for radio communications, regenerating the decompanding voice signal envelope would perhaps do fine, as shown above. However for broadcast and movie sound track, the voice envelope signal may perhaps be needed to to sent along with the compressed signal, so the sound envelope is also relayed to the receiver to truely recover the compressed audio back into its original pre transmitted form, by using the original sound envelope control signal to recover the compressed signal.

As the circuit is not acting a constant volume amplifier, a noise gate may perhaps not be required, but only testing would tell if so a noise gate is needed.

Now by watching a YouTube video relating DSP filters for ham radio audio filtering, one YouTube video demonstrated the "Timeline DSP-9" unit. The filtering performance was to say the least impressive, so I thought how could this be achieved from analogue circuits.

By in large, FM stereo car radios sometimes came with a nice chip that would vary the stereo separation of the FM stereo audio signal according to the RF strength of the received FM stereo radio signal. I have tried this principle before but this was many years ago, perhaps I should search out this chip and have another crack at it.

However, consider the circuit below:

Now this above circuit is perhaps a we bit interesting. Now lets assume that the Parametric filter ( the gyrator BPF ) is tuned or set to 500Hz, on the principle that your incoming HF signal audio may or not have a 500Hz audio spot point within the recovered audio on your HF radio set speaker. If the incoming audio spectrum width has a 500Hz base line spot frequency component, then the above circuit will allow that part of the voice signal to pass through. If however, the voice signal is of a higher pitch, there may not be a 500Hz component, and so the background noise of the audio around the low portion of the voice will be attenuated. Likewise, if the spot component signal spot is say 2KHz, the same would happen here. Should the 2KHz spot be within the audio, it would pass through, but if not, the 2KHz spot would be attenuated from the speaker audio, thus the back ground hiss signal around the 2KHz point.

Actually the above circuit is in error. To bias the Jfet , use the next circuit above this circuit to correctly bias controll the jfet operation principle. The next circuit above this circuit, the "Jfet plus R1" accumulate to the input resistance to control the op-amp overall gain. Should the voice signal volume be high, the JFet will increase in resistance, reducing the op-amp signal gain. However if the voice signal is low, the JFet would have its resistance reduced, thus increasing the op-amp overall gain. With this arrangement, a constant volume amplifier would be created, and if the gyrator bandpass filter is placed on the circuit input, then a standard level of bandpass signal response audio would be present upon the microphone radio audio signal.

An alternative setup would be use the above principle to max limit the bandpass filter response audio on any particular voice audio, from either the microphone into the radio, or the recovered audio from the radio itself.

In essence, three types of audio processor coule be craeted: 1/ a standard microphone parametric equaliser and limiter module, 2/ type (1 ) as also as a bandpass max audio response module, or 3/ a type ( 1 ) module as also as a constant level bandpass module.

In short, if there were say five tuned parametric dynamic audio processor circuits, set across the audio spectrum for voice or perhaps a HiFi audio for a greater number of tuned parametric processor circuits, each output of each one connected to a multi input audio mixer around an op-amp, the audio signal bandpass would then be depended upon the audio spectrum content of the received audio signal.

If the audio quality is short, otherwise narrow in voice bandwidth, then the above filter tuned section would match the received narrow audio quality. If on the other hand the receive voice quality is good, then the dynamic parametric filter would allow through the whole voice audio signal, to the spectrum bandwidth as received.

The principle being to design a filter bandpass circuit that is adjustable to the quality if the received audio signal. Hopefully the background hiss from todays radio speaker performance, would only be that of the received audio signal bandwidth, as the parametric dynamic audio processor would adjust itself to the bandwidth of the received audio voice signal.

The alternative to using a jfet as a variable attenuator, the audio compressor / expander chip refrred to as the "NE570", could be used instead. The pin connection to the agc control circuit, is best replaced by a purpose made audio agc circuit voltage. If the mic audio is just shaered between the agc generator and the mic amp, on the ne570, then all that is acheived is the mic audio reducing in volume.

By creating one own agc voltage o feed into the ne570 agc input, the spikes of the voice signal can then only be reduced, the threshold of the compression then variable. Atfer all, it is the audio peeks or spikes that are needed to be reduced, on order not to top and tail the transmitter signal output.

The NE570 has two halfes, in being that one half can be used for the TX, while the second half used for RX audio impluse suppression, but also limiting the dynamic range of the audio peeks or spikes from distorting the audio stage fo the RX radio section.

Similarly, the NE570 chip can be used in the circuit idaes shown here, i.e. for the paramteric equaliser application, one chip compressor for each bandpass filter fo the parametric equaliser.

An alternative use for the Volume unit meter, ( Vu meter ), an idea I had thought over, and many radio hams have also derived the same idea, and that is to use one Vu meter for the microphone connection, feed the the voice signal through the parametric equaliser and compressor limiter, the NE570 compressor chip, then onto a second Vu meter connection as an voice processor circuit signal output into the microphone radio input connection.

The first Vu meter would show the natural voice peeks, the second Vu meter the processed voice into the radio, and hence then the controlled voice level peek signals, that would otherwise over drive the RF final stage amplifier, and also prevent over driving the RF power amplifier.

The other manner to avoid over driving the radio, is to keep a watch on the RF signal power output meter on the radio, and use the voice from the microphone human volume to prevent bouncing the transmitter off the ceiling of the RF final stage, and the RF power amplifier.